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We propose and demonstrate the first RF digitally controlled oscillator (DCO) for cellular mobile phones. The DCO is part of a single-chip quad-band fully compliant GSM transceiver realized in a 90 nm digital CMOS process. Wide and precise linear frequency tuning is achieved through digital control of a large array of standard n-poly/n-well MOSCAP devices that operate in flat regions of their C-V curves. The varactors are partitioned into binary-weighted and unit-weighted banks that are sequentially activated during frequency locking and tracking. The finest varactor step size is 12 kHz at the 1.6-2.0 GHz high-band output. To attenuate the quantization noise to below the natural oscillator phase noise, the varactors undergo high-speed second-order 61 dithering. We analyze the effect of the 61 dithering on the phase noise and show that it can be made sufficiently small. The measured phase noise at 20 MHz offset in the GSM900 band is 165 dBc/Hz and shows no degradation due to the 61 dithering. The 3.6 GHz DCO core consumes 18.0 mA from a 1.4 V supply and has a very wide tuning range of 900 MHz to support the quad-band operation.

Texas Instruments, Dallas, TXThe use of deep-submicron CMOS processes allows for an unprecedented degree of scaling in digital circuitry, but complicates implementation and integration of traditional RF and analog circuits. The explosive growth of cellular radios makes it imperative to find digital architectural solutions to these integration problems. A fully-digital frequency synthesizer and GFSK transmitter for a single-chip Bluetooth radio is proposed in [1]. In this paper, a second generation of the digital radio processor (DRP) targeting GSM/EDGE cellular radios is presented. The alldigital PLL (ADPLL) phase-noise performance is significantly improved through architectural and circuit enhancements, and its wideband frequency modulation capability is extended to accommodate wide frequency deviations for EDGE. To complete the polar TX modulation path, a fully digital amplitude modulation circuit is added.At the heart of the ADPLL (Fig. 17.5.1) lies a digitally-controlled oscillator (DCO) [2]. The oscillator core (Fig. 17.5.2) operates at twice the 1.6 to 2.0GHz high-band frequency, which is then divided for precise generation of RX quadrature signals. The single DCO is shared between the TX and RX, and is used for both the high bands (HB) and the low bands (LB). Additional 4b of the tracking bank are dedicated for ∆Σ dithering in order to improve frequency resolution. The ADPLL sequencer traverses through the PVT calibration and acquisition modes during channel selection and frequency locking and stays in the tracking mode during the transmission or reception of a burst. To extend the DCO range to accommodate for voltage and temperature drifts, and to allow wide frequency modulation, the coarser-step acquisition bits are engaged by subtracting an equivalent number (generally fractional) of the tracking bank varactors. The acquisition/tracking varactor frequency step calibration is performed in the background with minimal overhead using dedicated hardware. All the varactors are realized as n-poly/n-well MOSCAP devices that operate in the flat regions of their C-V curves. The new varactors and the DCO core structure result in better phase noise than in [2], which is needed to meet the stricter GSM requirements.The ADPLL operates in the phase domain as follows: The variable phase of the ADPLL is digitally represented by a fixed-point concatenation of the DCO edge-transition count R V [k] and the normalized time-to-digital converter (TDC) output ε[k]. The TDC measures and quantizes the time differences between the frequency reference (FREF) and the DCO edges. The sampled differentiated variable phase is subtracted from the frequency command word (FCW) by the digital frequency detector. The frequency error f E [k] samples are accumulated to create the phase error φ E [k] samples, which are then filtered by a fourth-order IIR filter and normalized by a proportional loop attenuator α. A parallel feed with coefficient ρ adds an integrated term to create type-II loop characteristics, which suppresses the DCO flicker n...

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This paper addresses the design and properties of an intelligent optimal control for a nonlinear flexible robot arm that is driven by a permanent-magnet synchronous servo motor. First, the dynamic model of a flexible robot arm system with a tip mass is introduced. When the tip mass of the flexible robot arm is a rigid body, not only bending vibration but also torsional vibration are occurred. In this paper, the vibration states of the nonlinear system are assumed to be unmeasurable, i.e., only the actuator position can be acquired to feed into a suitable control system for stabilizing the vibration states indirectly. Then, an intelligent optimal control system is proposed to control the motor-mechanism coupling system for periodic motion. In the intelligent optimal control system a fuzzy neural network controller is used to learn a nonlinear function in the optimal control law, and a robust controller is designed to compensate the approximation error. Moreover, a simple adaptive algorithm is proposed to adjust the uncertain bound in the robust controller avoiding the chattering phenomena. The control laws of the intelligent optimal control system are derived in the sense of optimal control technique and Lyapunov stability analysis, so that system-tracking stability can be guaranteed in the closed-loop system. In addition, numerical simulation and experimental results are given to verify the effectiveness of the proposed control scheme. Index Terms-Flexible robot arm, fuzzy neural network (FNN), intelligent control, optimal control, permanent-magnet (PM) synchronous servo motor.

Normal ECG values in newborns, infants, and children have been collected and published. ECG in the adolescent, however, remains, to be collected and studied. The present study was designed and carried out to establish the normal ECG standards in male and female adolescents. A total of 898 school children and adolescents screened and examined as healthy were divided by age and sex into 6-9, 9-13, and 13-18 years age-groups. A 12 lead conventional ECG was recorded in 10 mm/mV and 25 mm/s, utilizing an automated Fukuda Denshi FCP-4301, MS-DOS/IBM-AT ECG machine. Lead V3R was not taken. Analog-to-digital conversion was performed by Fukuda signal acquisition module at a sampling rate of 500 Hz. The data on 69 ECG parameters were analyzed for the mean, standard deviation, 2nd to 98th percentiles, 95% confidence intervals, and sex difference. Normal values on 69 ECG parameters, sex-specific heart rate, P-QRS-T interval, duration, axis, wave amplitude, and calculated R/S amplitude ratio and ventricular activation time by age-group and sex were established. Male and female difference was noted in 49 (71.0%) parameters, of which 3 (6.1%) began in 6-9 years age-group, 30 (61.2%) began in 9-13 years age-group, and 16 (32.7%) in 13-18 years age-group. No sex difference occurred in 20 (29.0%) parameters. Normal male and female ECG standards on 69 ECG parameters in the adolescent were established. ECG sex difference began to appear the earliest at ages 6-9 years, and it occurred mostly at ages 9-13 years and 13-18 years, reflecting the anatomical and physiological consequences of puberty.