There has been an increased demand for 3G cell phones that support multiple bands of operation and are backward compatible with the 2G/2.5G standard to provide coverage where 3G networks have not yet been fully deployed. The transceiver design for such a handset becomes complicated with the need for separate transceivers for 3G and 2G/2.5G [1,2] or for multiple inter-stage receive / transmit SAW filters [3]. A single-chip transceiver that operates as a multimode multiband radio and eliminates the inter-stage receive / transmit SAW filters is presented. Figure 6.3.1 shows the block diagram of the transceiver with 7 primary and 4 diversity bands in WCDMA, and quad band in GSM. The transceiver is designed to operate in any of the UTRA bands 1 to 10, with the exception of band 7. It supports HSDPA (Cat 1-12), HSUPA (Cat 1-6), EGPRS (Classes 1-12, 30-39), and compressed mode of EGPRS / WCDMA operation. The transceiver is compliant with 3G DigRF interface 3.09.The analog / RF section of the transmitter is shown in Fig. 6.3.2. The I/Q DAC is based on oversampled current steering with 10b accuracy realized with a 6b-unary and 4b-binary segmentation. The output of the DAC is fed to a 3 rdorder Chebyshev continuous-time filter that attenuates out-of-band noise and DAC images and drives the I/Q modulator. The filter can be configured to operate in WCDMA, GMSK or EDGE mode. The I/Q modulator is a passive LO-2LO mixer. Two levels of passive switches are driven by LO and 2×LO frequencies with proper phases. This mixer configuration achieves both lower phase noise in the LO path and isolation between I and Q baseband inputs, and addresses the stringent linearity and noise requirements of GSM/WCDMA. The differential output of the I/Q modulator is converted to single-ended output using an on-chip balun. The balun output is then amplified in the driver stage and sent off-chip to the PA. The transmitter provides 80dB of gain-control range in WCDMA mode and 40dB in EDGE mode. The gain control is distributed across the chain and designed to meet the linearity and noise requirements over power-control range, while optimizing the current consumption. At high power (24 to 0dBm) a closed-loop power control scheme in the transceiver is used for accurate power control, and at lower power (0 to -57dBm) a conventional power-control scheme through the base-station is used.The architecture of the receiver front-end is shown in Fig. 6.3.3. The mixers and LO buffers are shared among several LNAs to reduce die area. The mixer outputs are connected to the virtual grounds created by the transimpedance amplifiers (TIAs) and its inputs are driven by the output current from the LNA. In this current-driven passive-mixer topology, the voltage swings at the mixer input and output are significantly reduced, resulting in improved linearity. In a passive mixer design, the noise contribution of the TIA increases as the impedance at the mixer input decreases. Therefore, an LC tank is used at the LNA cascode output with switchable capacitors to adjust the tan...

-This paper presents an integrated wide-range VCO with a modified tuning scheme to deal with VCO frequency drift over temperature. In this approach, during the coarse-tune operation, VCO tune voltage is a function of temperature such that it resembles the inverse function of VCO fine-tune characteristic. Without degrading VCO performance, the proposed temperature adaptive tuning optimizes the maximum tolerable VCO temperature frequency drift over which PLL remains locked. As a result, VCO gain can be reduced significantly, making VCO less sensitive to PLL tune voltage noise. Integrated in a multistandard multi-band transceiver with a small VCO gain of 50MHz/V at 3.90GHz, PLL remains locked despite 45MHz frequency drift of VCO over [-30ºC, 85ºC]. Using an on-chip inductor, VCO covers from 3.15GHz to 4.60GHz, achieving -138.0dBc/Hz phase noise at 3.0MHz at 3.90GHz by drawing just 8.5mA from 1.60V supply in 0.13u CMOS process.Index Terms -Wide-range VCO, VCO temperature frequency drift, VCO fine-tune , on-chip inductor, PLL

Background: Publications have focused on comparing the effect of various methods of refractive surgery or ablation profiles on higher order aberrations (HOA), yet the effect of WG compared to TG ablation in femtosecond laser-assisted in-situ keratomileusis femto-LASIK on HOAs has not been studied adequately. Materials and methods: We reviewed the changes of HOAs before and after femto-LASIK for simple myopia or compound myopic astigmatism after topography-guided (TG) and wavefront-guided (WG) ablation profiles. In this retrospective study, 42 eyes underwent TG and 31 WG femto-LASIK. Results: The total HOAs increased from 0.41±0.16 μm pre-operatively to 0.66±0.30 μm post-operatively in the TG group and from 0.42±0.13 μm pre-operatively to 0.59±0.27 μm post-operatively in the WG group without statistically significant difference between the two groups. However, the postoperative change of secondary astigmatism showed a statistically significant difference between the two groups [0.20 (0.0-0.53) vs. 0.0 (0.0-0.20) for TG and WG groups, respectively; p=0.009]. Conclusions: The total HOAs increased compared to pre-operative data, in both the TG and WG ablation profile groups. The extent of increase of secondary astigmatism was significantly lower in the WG group, which might favour the postoperative quality of vision.

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