The dynamic range, required by wireless cell-phone terminals, makes the design of analog circuits at the supply voltage available in deep submicron CMOS technologies so challenging that researchers are investigating alternative methods to process the received signal [1]. On the other hand, several analog techniques, at RF in particular, have emerged in the recent past, improving both linearity and noise performances of different building blocks, due to an in-depth understanding of the underlying physical processes [2,3]. In this paper, a direct-conversion fully integrated RF front-end is proposed for GSM. Realized in 90nm CMOS, the chip operates at a supply voltage compatible with the 45nm node and demonstrates the following main performances: 3.5dB NF integrated between 1kHz and 100kHz, 15kHz 1/f noise corner, 51dBm minimum IIP2, at a supply voltage as low as 750mV. No calibration is required to meet the IIP2 specification.GSM is particularly demanding due to closely spaced channels and huge interferers surrounding the received signal. The resulting required 1/f noise corner and IIP2 usually suggests disregarding a direct-conversion architecture when the technology is CMOS, despite its attractiveness to arrive at a highly integrated transceiver implementation. The RF front-end can not rely on any frequency selectivity, having the most dangerous interferers only few MHz away from the received signal. As pointed out in Fig. 26.1.1, the down-converter is the most critical block where low 1/f noise and high linearity are conflicting requirements, at given current consumption [3,4]. Furthermore, the reduction in the available supply voltage of scaled processes calls for minimum device stacking, making a conventional Gilbert cell not attractive. In particular, a fully differential transconductor, though showing excellent IIP2 performances, determines a significant voltage drop to accommodate the biasing current source. The pseudo-differential alternative has the advantage of a lower voltage drop and a superior IIP3, though at the expense of a dramatically reduced IIP2. The second-order common-mode conductance gain is at the origin of much higher second-order inter-modulation products in the mixer output current [4]. The idea proposed in this paper is to minimize the inter-modulation current produced by a pseudo-differential transconductor by means of a feedback loop controlling the mixer output common-mode voltage. Referring to Fig. 26.1.2, where the mixer schematic is drawn, an opamp senses the difference between the common-mode output voltage and a reference and drives the PMOS device, P CM , supplying the common-mode current to the input transconductor. By inspection, the loop gain (G loop ) is given by:(1) where A is the opamp gain, g m,PCM is the tranconductance of P CM , and r ds,PL the output resistance of P L .The loop operates at a low frequency and can produce a large gain at minimum consumption, significantly reducing the commonmode inter-modulation current injected into the switching pair. To gain quantita...